The present invention relates in general to communication systems, and is particularly directed to a new and improved encoding and signal processing scheme for extending the normal range of digital communications transported over a two-wire telephone wireline channel, such as, but not limited to, a high speed data service loop (HDSL), a digital data service (DDS) channel and an integrated services digital network (ISDN) channel, to distances (e.g. on the order of 25 Kft) well beyond those currently possible (typically on the order of 15-18 Kft for a basic rate ISDN channel) using a repeater-less two-wire transmission path.
In order to meet various ANSI requirements for digital data communications, telephone subscriber copper wire lines must meet specified industry standard performance criteria, which limit the operational range of a two-wire loop. For example, in the case of currently installed ISDN basic rate digital subscriber lines (having a data rate of 160 kilobits per second, including bidirectional data payload and overhead maintenance channels), the ANSI standard T1.601 for 2B1Q (two-binary/one, quaternary/four level) modulation, two-wire, full-duplex data transport with echo cancellation, typically describes an ISDN channel as one that does not exceed a two-wire loop loss of 42 dB at 40 KHz, or 1300 ohms, resistive. As a consequence, the operational range of such a two-wire loop is limited to a range on the order of 15-18 kft, using No. 26 (American Wire Gauge) wire, and commercially available ISDN transceiver interface equipment.
To extend ISDN communications to the approximately twenty percent customer premises market that lies geographically beyond this range, it is necessary that the service provider either install repeaters in the loop, or use a different communication medium, such as a T1 carrier fiber optic link. Unfortunately, each of these alternative solutions to the extended range problem carries with it a substantial cost penalty that the customer is unwilling to bear.
For example, the repeater approach requires the installation of both an office end repeater powering unit, plus a repeater mounting pole, or a subterranean, environmentally hardened housing (bell jar) for the repeater. Not only does this involve the use of additional equipment (including the cost of the repeater hardware and its installation), but it entails the expense and labor of maintaining the repeater enclosure.
Similarly, although T1 channel banks, located in both the switch office and another downstream location (office or remote hut) that is geographically xe2x80x98closexe2x80x99 to the subscriber premises, are able to accept basic rate interface transmission extender (BRITE) cards for T1 carrier extension, the fact that T1 carrier systems are configured to include capacity for multiple extended basic rate services means that their use to deliver only a single basic rate extended service is prohibitively expensive and impractical.
In accordance with the present invention, the desire to extend the range of digital data communication services (such as a basic rate ISDN channel) to customer premises located beyond the presently achievable two-wire loop range (e.g. on the order of 15.2 kft for basic rate ISDN lines), without the above described cost penalty, is successfully addressed by: 1xe2x80x94changing the line code or modulation format; and 2xe2x80x94adopting enhanced signal processing techniques, which may be of the type employed in high bit rate digital subscriber line (HDSL) systems, to accommodate a diminished signal-to-noise ratio (resulting from the added insertion loss inherent in the extended length of the two-wire pair).
Considering the application of the present invention to the case of an ISDN channel, as a non-limiting example, advantage is taken of the availability of what have now become reasonably priced integrated circuit-based signal processing components, such as high speed digital application specific integrated circuit chips (digital ASICs), whose processing power and speed greatly reduces the cost of implementing a relatively sophisticated digital communication transceiver, particularly one that is intended to operate at data rates considerably reduced compared to the high speed data processing capacity of digital ASICs.
With the availability of these cheaper components, the overall cost of incorporating low signal-to-noise ratio signal processing techniques into a slower data rate transceiver, such as, but not limited to an ISDN device operating at only one-fifth the data rate of an HDSL scheme, is far less than that required to implement: either of the conventional range extension approaches, described above, such as that involving the installation of a repeater.
Pursuant to a first aspect of the present invention applied to ISDN communications, the symbol rate of customarily employed 2B1Q ISDN line code modulation scheme for a basic rate ISDN channel is modified via an encoding and translation operator which achieves a reduction in symbol rate equivalent to transmitting three information or payload bits per symbol (a construct for which may be expressed as or represented by a 3B1O (three binary, one octal/eight level) line code), instead of the two bits per symbol that are transmitted using 2B1Q line code modulation. For a 160 kilobits per second ISDN basic rate interface, this initial symbol rate reduction of transmitting three information or payload bits per symbol instead of two bits per symbol means that the same number of information bits can be transmitted at two-thirds the standard symbol rate, or at a symbol rate of 53,333 symbols per second, which has the inherent property of increasing the transmission distance over the two-wire link that will comply with the above-referenced ANSI loss standards.
For this purpose, in the environment of a full-duplex data communication system, employing echo cancellation, with a transceiver (transmitter and receiver) installed at each end of the data transport link, the front end of a transmit section of a transceiver configured in accordance with the invention is coupled with a standard, basic rate ISDN transceiver interface U-chip which receives a basic rate (80 kilobits per second) ISDN signalling channel carrying 2B1Q signals (such as those sourced from a central office for transmission to a customer premises site, or sourced from a customer premises site to the central office). The U-chip is clocked so that it outputs 2B+D formatted digital signals (and overhead signals) to a framing unit, which assembles the digital signals and any accompanying overhead bits into a serial framing format, and outputs the respective bits of the serial frame to a serial data scrambler. The data scrambler randomizes the data so as to ensure full spectral occupancy of the transmission band on the transport link, enabling proper operation of adaptive elements in the receiver.
The scrambled serial data stream is converted into a three-bit parallel format by a serial-to-parallel converter and coupled to an error correction encoder, such as a trellis encoder, which forms part of the above-mentioned enhanced signal processing mechanism of the second aspect of the invention. The trellis encoder may introduce redundancy causing a rate of 4/3, so that it produces a four bit code from each group of three information bits. The trellis-encoded four bits are then translated or mapped via a code translator into a 4B1H (four bits, one hex/sixteen level) output line codexe2x80x94representative of one of sixteen levels of a pulse amplitude modulated signal to be transmitted per symbol.
As a non-limiting example, and maintaining compatibility with telephone industry standard parameters, the code translator may employ a sixteen-valued, one-dimensional linear signal space comprised of data points having relative values of: xe2x88x9215, xe2x88x9213, xe2x88x9211, xe2x88x929, xe2x88x927, xe2x88x925, xe2x88x923, xe2x88x921, +1, +3, +5, +7, +9, +11, +13, +15. Thus, for each symbol, the 4B1H line code translator will output a respective four bit line code representative of one of the sixteen data values, as determined by the trellis-encoded value.
In order to accommodate a normally unacceptable reduction in signal-to-noise ratio that would inherently result from the insertion loss due to increased length of the two-wire loop, the second aspect of the invention further uses reduced signal-to-noise ratio signal processing techniques, such as Tomlinson precoding and adaptive equalization that are readily implemented using the available digital signal processing power of the previously mentioned relatively inexpensive, digital ASICS.
Such enhanced signal processing may further include coupling the 4B1H code translator to a Tomlinson precoder, which is operative to weight the PAM encoded signal in accordance with a prescribed filtering operator, which is equivalent to the feedback filter that would be used in a decision feedback equalizer located in the receiver section of the peer transceiver if Tomlinson preceding were not being used. Because the data has been trellis-encoded, however, a decision feedback equalizer cannot be used in the downstream receiver, since the two are mutually incompatible. This incompatibility problem is remedied by the use of the Tomlinson precoder in the transmitter, which achieves the same performance in the presence of intersymbol interference and noise as the use of a decision feedback equalizer in the receiver.
The Tomlinson precoder includes an adder which sums the four bit 4B1H code generated by the code mapping translator with the output of a multitap filter. The multitap filter may include an finite impulse response (FIR) filter and an optional infinite impulse response (IIR) filter, and has its weighting coefficients established during a training mode of operation, in which an adaptive equalizer in the receiver section is configured and operated as a decision feedback device. The multitap filter is coupled to receive the output of a modulo index operator unit in the Tomlinson precoder. The modulo index operator, termed a modulo unit, is operative to adjust the output of the adder, as necessary, based upon integral multiples of the magnitude of the range of the PAM coding space, such that the output of the unit falls within the coding range of the PAM signal (between the values: xe2x88x9216 and +16, for the above 4B1H line code example).
Because of the effect of the multitap filter, the value supplied to modulo unit will customarily be a non integer, whereby the output of the Tomlinson precoder has an effectively continuous signal characteristic. This effectively continuous, multi-bit, Tomlinson-precoded signal is applied through the combination of a digital-to-analog converter and low pass transmit shaping filter for transmission over the two-wire telephone channel by way of a line coupling circuit, such as a transformer interface unit.
In the receiver section, the two-wire loop is terminated via a line coupling circuit, the output of which is digitized by an analog-to-digital converter. The analog-to-digital converter is clocked by means of a recovered clock signal derived from a timing recovery circuit.
To remove the effects of transmit echo introduced by the transmit section of the local transceiver, the output of the analog-to-digital converter is differentially combined with the output of an adaptive echo replica filter in an echo-canceling unit. The adaptive echo replica filter is coupled to the output of the Tomlinson precoder in the transmit section and outputs a replica of the echo signal that is coupled from the transmit section through the line-coupling circuit to the local receiver.
The output of the echo canceler is coupled to an adaptive equalizer which, during normal operation is configured and operated as an adaptive linear equalizer. During a training mode, the device is configured and operated as a decision feedback device for the purpose of established weighting coefficients to be applied to the filter structure of the Tomlinson precoder at the far end of the link. During this training mode, the above-referenced error-correcting trellis encoder within the transmit section of the transceiver at the other end of the link is disabled.
Once the weighting coefficients of the decision feedback equalizer have adapted to steady state values, the coefficients are forwarded to the peer transmitter, and are used to set the tap values of the multitap filter of its Tomlinson precoder.
With the tap values of the Tomlinson precoder filter within the transceiver at the far end of the link set for normal operation, that transceiver""s Tomlinson precoder filter is functionally equivalent to having installed a decision feedback equalizer in the receiver, but circumvents the incompatibility problem described previously. Upon completion of the training mode, the trellis encoder is enabled, and the decision feedback section of the adaptive equalizer of the receiver is disabled, so that the taps of the adaptive equalizer may operate in its normal mode.
The output of the adaptive equalizer is coupled to a further modulo unit which, like the modulo unit in the Tomlinson precoder of the transmit section, is operative to add or subtract an integral multiple (m) of the modulo value (i.e. mxc3x9732 in the present example of a dual polarity range from xe2x88x9216 to +16) from the output of adaptive equalizer, where (m) is a positive or negative integral multiple, so that the output of modulo unit in the receiver section will fall within the range of the one-dimensional signal coding space of the PAM signal. The output of this modulo unit is coupled to a Viterbi decoder, which decodes the modulo-translated signal value into a three bit parallel word, corresponding to the original group of three information bits input to the trellis encoder of the transmit section of the transceiver at the far end of the link.
The recovered three bit parallel data output by the Viterbi decoder is then converted into serial format by a parallel-to-serial converter, so that it may be descrambled by a serial descrambled which outputs the original successive data frames to a framing disassembly unit. The framing disassembly unit disassembles the frame-formatted digital signals into respective overhead signals and the original 2B+D signals. The 2B+D signals are coupled to an attendant U-chip, which outputs the original 2B1Q-formatted basic rate ISDN channel for delivery to an attendant destination device, such as customer premises terminal equipment.